Multifunction receiver-on-chip for electronic warfare applications

ABSTRACT

What is provided is a receiver-on-a-chip comprising a monolithic integrated circuit that reduces the receiver to a cigarette-pack-sized assembly mountable directly at an antenna element, with a much-increased operational bandwidth and instantaneous bandwidth, increased dynamic range and with a two-order-of-magnitude decrease in size and weight. Moreover, because of the elimination of all of the I/O drivers and attendant circuitry, power consumption is reduced by two-thirds, whereas the mean time before failure is increased to 10,000 hours due to the robustness of the monolithic integrated circuit and use of fiber optics.

RELATED APPLICATIONS

This is a divisional of patent application Ser. No. 12/386,681 filedApr. 22, 2009 and patent application Ser. No. 10/543,343 filed Jul. 22,2005 which issued as Pat. No. 7,542,812 and issued on Jun. 2, 2009entitled Multifunction Receiver-On-Chip For Electronic WarfareApplications, the contents of which are incorporated herein byreference.

STATEMENT OF GOVERNMENT INTEREST

The invention described herein was made under Contract No.DAAB07-02-C-K513 with the Government of the United States of America andmay be manufactured and used by and for the Government of the UnitedStates of America for Governmental purposes without the payment of anyroyalties thereon or therefor.

FIELD OF THE INVENTION

This invention relates to receivers used in electronic warfare and moreparticularly to a miniaturized receiver-on-a-chip, placeable directly atan antenna element in which prior electronic warfare, EW, receivermodules are integrated monolithically into a single integrated circuitusing ultra high-speed transistors.

BACKGROUND OF THE INVENTION

For many electronic warfare applications, phased array antennas are usedfor beam forming or direction finding purposes in which an aircraft, forinstance, is provided with an antenna array, the outputs of the elementsof which are coaxially cabled to an equipment bay on the aircraft wherethe signals are processed through a number of different receivermodules. These modules include low noise amplifiers, analog-to-digitalconverters, filters, mixers, IF stages, amplifiers and processors thatconstitute a centralized receiver system that is complex, heavy,consumes an excessive amount of power and is expensive.

In addition to weighing in excess of 60 pounds, the input/output (I/O)drivers of such a modular system can consume as much as 15 watts ofpower out of a 25-watt total, with the power drain primarily residing inthe interfaces between the modules. This is because the interfaces mustemploy drivers that consume an excessive amount of power. Moreover,cabling between the modules and to antenna elements is heavy and leadsto cable losses that result in power drain and decreased receiversensitivity.

Power consumption and weight are indeed factors when one seeks toprovide an unmanned aerial vehicle (UAV) such as the Predator withelectronic warfare (EW) receiver capabilities. It will be appreciatedthat unmanned vehicles have limited fuel supplies, or if powered bysolar cells, can only accommodate equipment having very limited powerconsumption. When the UAVs hover over an area sometimes for days, weeksor months, the longevity of the mission is critically dependent uponfuel consumption, which is in turn directly related to power consumptionof the avionics package. Moreover, the ability to reduce the weight fromthe 60-pound modular system described above is critical because weightreduction translates to increased endurance. Size reduction is also afactor because present rack-mounted modular EW systems occupy too muchspace to be incorporated into the UAV avionics package.

It will thus be appreciated that in high-altitude, long-endurance UAVsthe physical size of the avionics package is a problem. Not only doespower consumption translate into endurance, but the ability to do thesignal processing associated with the EW receivers must be done inpackages that are to be located on a platform that is ten times smallerthan, for instance the P3 reconnaissance aircraft.

In short, if one were to be able to completely eliminate the modules andthe extensive coaxial cabling between modules, one could significantlyreduce size, weight and power consumption, while at the same timereducing impedance mismatches that reduce sensitivity.

While one might be inclined to produce an EW receiver using multi-chipmodules or MCM technology, it will be appreciated that the multi-chipmodule approach also consumes a significant amount of power. While themulti-chip module can shrink the size of the system to a certain extent,one must address the I/O interface power requirements, which asmentioned above can result in 15 watts wasted power out of the 25-watttotal requirement. Thus, for instance, if one were to make a modularreceiver system having, for instance, one module that includes ananalog-to-digital converter and a demultiplexer, a second module thatcontains a low band pass filter, a high band converter, clock and localoscillator generation, a digital automatic gain control coupled to ananalog-to-digital converter and another demultiplexer, all of which arecoupled to a CMOS DSP processor, which is in turn coupled to aserializer, one would expend 5 watts of I/O power associated with thefirst analog-to-digital converter. This power consumption is added to a2-watt current consumption for the I/O to the CMOS digital DSP. Next,there is a loss of 4 watts of power for the output due to the I/Oassociated with the demultiplexer that is associated with the low bandand high band converters, with another 2 watts associated with the I/Oto the CMOS DSP. Between the DSP and the serializer, there is another 2watts of lost power due to the I/O drivers, with another 1 watt of lostpower associated with the demultiplexer ahead of the serializer.

While power consumption of a modular receiver utilizing MCM technologyis indeed a problem, there is also a requirement to improve on all ofthe characteristics of an MCM system to not only decrease powerconsumption for increasing endurance, but also to increase the dynamicrange, provide improved instantaneous bandwidth, increase the operatingfrequency bandwidth beyond the usual 2 GHz to 18 GHz bandwidth, andincrease the mean time before failure (MTFB). Note further that oneneeds to be able to decrease the equipment size from the present size of200 cubic inches down to something considerably more manageable.

SUMMARY OF INVENTION

Rather than using either the MCM approach or any other modular approach,assuming for instance 5 modules, each six inches square, by usingsystem-on-a-chip (SOC) architecture and the new IBM silicon-germaniumtransistor technology with switch speeds in excess of 100 GHz one canreduce the entire EW receiver to a package the size of 3 inches by 2inches.

Moreover, since the interconnections between the various circuits thatare monolithically formed on the chip are on the order of microns inlength rather than wires of an inch or more, transmission line lossesare eliminated along with the weight associated with the coaxial cableinterconnects. Not only is the physical size reduced by an order ofmagnitude, but also because of the monolithic integrated approach, onecompletely eliminates the I/O buffers between the circuits, whichsignificantly reduces the power consumption.

Even more significant is the fact that having achieved areceiver-on-a-chip package size of 2 inches by 3 inches, each receiverin and of itself can be placed adjacent an antenna element, withdownstream communication being accomplished through fiber optic cablessuch that weight concerns, losses and impedance mismatches associatedwith coaxial cabling between the antenna elements and the processors arecompletely eliminated.

The ability to locate the EW receiver directly at the antenna elementprovides increased flexibility for the type of processing that can beachieved and greatly increases sensitivity by minimizing losses.

What makes possible the formation of such monolithic system-on-a-chipcircuits is the advancement in semiconductor technology by IBM thatinvolves new silicon-germanium techniques. Because of this new, fastertechnology the resulting transistor speeds and densities make thesystem-on-a-chip configurations possible.

For instance, recent IBM developments provide FTX-MAXes that relate tothe toggling frequencies of the transistors well over 100 GHz, asopposed to the present state-of-the-art FTX-MAXes of 20 to 30 GHz. Whatthis means is that there is an order of 3 to 4 times improvement in theswitch speed, which translates into improved analog-to-digitalconverters that do not require as much down-conversion as was heretoforethought required.

The result of being able to run the analog-to-digital converters at muchhigher rates increases the overall receiver bandwidth from the original2 GHz-to-18 GHz to 0.03 GHz-to-18 GHz. Moreover, the instantaneousbandwidth goes from 500 MHz to 2000 MHz, with the dynamic range beingimproved from 50 dB to 60 dB. The size when compared to the 200 cubicinches for the modular units goes down to 7 cubic inches, whereas aweight of ten pounds for MCM EW receivers goes down to 1 pound.Moreover, the Mean Time Before Failure increases from 1,000 hours to10,000 hours.

The overall impact of the use of a silicon-germanium system-on-a-chipapproach reduces the number of multiple IF down-conversion stages andnot only eliminates the requirement of a separate RAIU plus a separateRF unit, but also permits moving the analog-to-digital converter afterthe second local oscillator to the front end of the system. What thismeans is that there is a parts count reduction due to the ability toeliminate two local oscillators and the associated down-conversions,simply by being able to design a super-high-speed analog-to-digitalconverter. Also gone are the drivers between modules and the attendantpower consumption.

Importantly, the heavy coaxial cables used to interconnect the modulesand, indeed, to connect the equipment bay to the phased array antennaelements are completely eliminated. Because the receiver has now beenreduced to a size that can be conveniently placed at each of the antennaelements, no coaxial cabling is required with its attendant losses andweight. The result is that one can do whatever signal processing isnecessary directly at each of the antenna elements and connect thedigital receiver outputs by fiber optic cables that are in essencelossless devices.

One therefore has gone from a centralized receiver system that iscomplex, heavy, power-consumptive and expensive to a much lower cost,more flexible system that can be placed at each antenna element. Theresult is that such a system can be easily deployed on UAVs and, forinstance, can be used on ground sensors that need to be low power andvery small because they are battery powered and must last for longperiods of time after deployment.

Gone also are the high-speed interfaces, which are power-consumptive,with the subject receiver-on-a-chip completely re-capturing the 15 wattslost in multiple module systems. Thus, the subject system is uniquelyapplicable to high altitude long endurance UAVs, which may have tocircle and loiter for weeks or months while performing persistent ISR orintelligence surveillance and reconnaissance.

As will be appreciated, when one goes from a 60-pound equipment baypayload down to 1 pound, endurance is optimized.

Moreover, the subject system achieves added flexibility since eachantenna element can now be provided with its own receiver and its owntuner. One can therefore do much more flexible digital beam forming withthe multi-function operation achievable by the subjectreceiver-on-a-chip. As a result, anything that had required analogsignals can now be put efficiently digitized and put together into asingle receiver channel. Moreover, since one does not have to deal withthe losses associated with cabling, one achieves better receiversensitivity.

It has been found that one can achieve an 11 dB increase in sensitivityby eliminating the coaxial cabling. And one can also achieve a bandwidthincrease below 2 GHz. As mentioned above, because of the higher-speedanalog-to-digital converter capability afforded by the newsilicon-germanium technology, one can have a 10 GHz-per-secondanalog-to-digital converter as compared with the present 2 GHzanalog-to-digital converters. The increased speed of theanalog-to-digital converters directly translates to increased bandwidthand reduces the number of IF stages.

In summary, what is provided is a receiver-on-a-chip comprising amonolithic integrated circuit that reduces the receiver to acigarette-pack-sized assembly mountable directly at an antenna element,with a much-increased operational bandwidth and instantaneous bandwidth,increased dynamic range and with a two-order-of-magnitude decrease insize and weight. Moreover, because of the elimination of all of the I/Odrivers and attendant circuitry, power consumption is reduced bytwo-thirds, whereas the mean time before failure is increased to 10,000hours due to the robustness of the monolithic integrated circuit and useof fiber optics.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features of the subject invention will be betterunderstood in connection with a Detailed Description, in conjunctionwith the Drawings, of which:

FIG. 1 is a diagrammatic illustration of the reduction in size for amodular EW receiver having a number of individual modules to amonolithic integrated circuit receiver having dimensions of 3″ by 2″,which can be located directly at an antenna element embedded in the wingof a fighter aircraft;

FIG. 2 is a diagrammatic illustration of the placement of the subjectmonolithic integrated circuit receiver at one of the phased arrayelements of an antenna carried by an aircraft, showing an 11 dB increasein sensitivity from reduced cable loss;

FIG. 3 is a diagrammatic illustration of the formation of a multi-beam,multi-function element level digital beam former array capable of beingembedded in the wing of the aircraft of FIG. 2;

FIG. 4 is a diagrammatic illustration of the use of the subject receiverin an expendable, intelligent, high-performance sensor, which isdeployable by air and which is battery powered, with the subjectreceiver-on-a-chip responsible for increased endurance;

FIG. 5 is a block diagram of the subject receiver-on-a-chip,illustrating that the entire circuit can be provided as a singlemonolithically formed silicon-germanium integrated circuit;

FIG. 6 is a block diagram of a prior art broadband EW receiver system,indicating that the individual multi-chip modules that togetherconstitute the receiver are provided with drivers for interfacing themodules, the use of which results in excessive power drain;

FIG. 7 is a block diagram illustrating the reduction in parts countassociated with taking a multi-chip EW receiver, in which the singleintegrated circuit receiver permits eliminating 2 IF stages and uses theanalog-to-digital converter at the head end of the subject system;

FIG. 8 is a table listing the differences between a state-of-the-artmulti-chip module EW receiver and the subject receiver-on-a-chip,showing improvements in operating frequency, instantaneous bandwidth,dynamic range, size, weight and mean time before failure;

FIG. 9 is a top plan photograph of a fabricated 2 to 18 Hz LNA of thepresent invention;

FIG. 10 is a graph showing measured LNA 5-parameters and noise figure;

FIG. 11 is a top plan micrograph of a variable gain amplifier of thepresent invention;

FIG. 12 is a schematic diagram of the core of the variable gainamplifier of the present invention;

FIG. 13 is a graph showing measured UGA power curves;

FIG. 14 is a schematic block diagram of a ADC/Serializer of the presentinvention;

FIG. 15 consists of graphs showing the results of two tone input testwith and without dither;

FIG. 16 is a schematic diagram showing a 20 GPS's 3 bit ADC/DAC; and,

FIG. 17 is a graph showing a 9.17 GHz tone sampled at 20 Gsps Max spur−26 dB.

DETAILED DESCRIPTION

Referring to FIG. 1, current modular digital EW receivers have a numberof modules 10, 12, 14, 16 and 18, respectively an RF converter module, amodule for the first half of an IF conversion section, a second moduleof an IF conversion section, an RF digitizer module, and an RF digitalprocessing module. These modules are cabled together in an electronicssuite within an aircraft so as to provide the requisite digital receiverfunctions for surveillance, intelligence gathering and oftentimes fordirection finding and fire control. Typically, each of these modules is6 inches on a side so as to comprise at least 30 inches of rack spacewithin the electronics suite of the aircraft.

As mentioned hereinbefore, these modules are usually interconnected bycoaxial cables as well as being connected by coaxial cable to variousantenna elements in the antenna array, usually embedded in the wing 20of an aircraft 21.

As mentioned hereinbefore, the weight, power consumption, losses andmean time before failure problems need to be resolved, not only toimprove bandwidth and lower power consumption and weight, but also toprovide a package that is easily installed in a much smaller aircraftsuch as a UAV.

As can be seen by the picture of receiver 22, it is possible to reduceall of modules 10-16 to a single monolithic chip and to provide the chipin a housing that in one embodiment is only 3 inches by 2 inches. In sodoing, there is a total volume reduction of greater than an order ofmagnitude, which offers significant performance improvement and enablesinsertion on high-endurance UAVs.

As illustrated in FIG. 2, a wing 24 of an aircraft is provided with aphased antenna array, in this case located at wingtip 26, whichincludes, for each element of the phased array, a receiver that is ofsuch small size that it can be co-located with the antenna element. Oneof the important features of the subject invention is that by making theEW receiver so small and co-locating it with an antenna element, one canachieve an 11 dB increase in sensitivity because, as will be describedhereinafter, any communication that is necessary between the receiversis accomplished through fiber optic cables, which offer a 40 Gb/s seriallink.

The combination of the receiver adjacent each element allows side lobeand back lobe electronic warfare and electronic intelligencecapabilities, while at the same time resulting in dramatic reductions insize, weight and recurring cost.

Referring to FIG. 3, diagrammatically, the array of antennas embedded inwing 24 in one embodiment includes a number of antenna elements 32mounted to wing 24, which enables multi-beam, multi-function elementlevel digital beam forming for the array, thus to remove operationalconstraints and increases the degrees of freedom, while enabling hyperconcurrent and cross-mission operation.

Referring to FIG. 4, such miniaturized EW receivers have use not only inairborne applications but also in battery-operated ground sensors suchas illustrated at 36, which are deployable by air dropping or physicalplacement, and which include communications equipment that must stay onstation for days, months or years and not run down the on-boardbatteries. Even if the sensors are provided with solar cells, the amountof current draw must be minimized and the supplying of the subjectreceiver-on-a-chip, with its high speed silicon-germanium technology,provides the deployable sensors with affordable receivers that can beused, for instance, for ground surveillance and situational awareness,or can be used for communication relays and to deploy electronic attack.

Referring now to FIG. 5, the subject receiver-on-a-chip 40 employs asilicon-germanium technology that not only reduces the parts count overa modular system but also results in increased bandwidth and other ofthe advantages noted above.

The feed for the subject receiver-on-a-chip is a 30 MHz to 18 GHz RFinput 42, which is coupled to an RF pre-select and attenuation or notchfilter circuit 44, which is off-chip. The output of this circuit isamplified by a low noise amplifier LNA 46 and is either switched betweena low band pass filter 48 or a mixer 50 is used in a singledown-conversion stage, with the mixer being driven by either 8, 10 or 12GHz local oscillators on line 52 from a clock and/or local oscillator54. Either the low pass filtered output of LNA 46 or the down-convertedoutput of a mixer 50 is supplied by a switching circuit 56 to anoff-chip IF filter bank 58, which provides IF filtering either for the2-4 GHz band or the 6-8 GHz band. What will be seen is that either thereis no down-conversion or there is only a single step of down-conversion.This operation is made possible due to the high frequencies at which theanalog-to-digital converters can be run due to the silicon-germanium IBMtechnology. The output of the IF filter bank is applied to avariable-gain amplifier VGA 60 under the control of a digital automaticgain control circuit 62, which as will be seen can be identical to mixer50. This minimizes the number of different circuits that need to befabricated.

Thus, as an offshoot of the subject invention, it will be appreciatedthat the same section of the monolithic circuit used for the mixer canbe used without alteration as both an RF down-converter using IBM SiGe 7HP technology or a variable-gain amplifier, also using this sametechnology. In the down-conversion mode, the RF down-converter has ameasured gain of 6.7 dB, with an 11.2 dB noise figure (NF) and a 2 GHzbandwidth.

As a variable-gain amplifier, the same technology has a bandwidth fromDC to 10 GHz, with a −24 to 14.5-dB gain and a 54-dB spur-free dynamicrange (SFDR).

It will be noted that the low noise amplifier 46 likewise utilizes IBMSiGe 7 HP technology, with a measured 21 GHz bandwidth and a 4.6 to5.8-dB NF.

It is noted that the output of the variable-gain amplifier 60 is appliedto an analog-to-digital converter 64, which has as one of its inputs a10 Gsps clock. The output of the analog-to-digital converter is appliedto a demultiplexer 66 and thence to a CMOS digital signal processor DSP68, which has standard interfaces to enable efficient applicationdevelopment. One output of the CMOS DSP 68 controls the digital-analoggain control circuit 62 over line 70, whereas a signal over line 72controls the RF pre-select/attenuate/notch circuit 44.

As illustrated, the output of the CMOS DSP 68 is applied to a serializer70, the output of which is 80 GPBS serialized output data, which can betransmitted over a fiber optic link to a digital processor.

Note that analog-to-digital converter 64, the CMOS DSP 68 and theserializer can be implemented by an analog-to-digital converter (ADC)serializer using IBM SiGe 7 HP technology, with a 10 Gsps 4-bitcapability, a 50-dB Spurious Free Dynamic Range (SFDR), with theserializer being a 40 Gbps serializer.

Note that an analog-to-digital converter 74 is coupled to the RF,pre-select, attenuate and notch unit 44, which is used to detectinterference over a full spectrum and provides a steerable notch filterfunction. This analog-to-digital converter is implemented in oneembodiment with a silicon-germanium 8 HP technology and has a 20 Gsps3-bit characteristic with a 26-dB SFDR. Note that the output of thisanalog-to-digital converter is demultiplexed at 76 and is applied to theCMOS DSP 68.

What will be appreciated is that the silicon-germanium high-speedtechnology is used in all components, namely the LNA, the low noiseamplifier, the RF down-converter, the variable-gain amplifier, bothanalog-to-digital converters, and for the serializer, with the CMOS DSPbeing the only element that employs CMOS technology.

As illustrated by reference character 80, the entire monolithicallyformed receiver-on-a-chip circuit may take the miniaturized form shown,which results in an IBM silicon-germanium 8 HP unit, with 10 Gsps at 6bits, a 60-dB SFDR, and can accommodate a CMOS DSP application having upto 1000 gates, with a 40 GPS serializer being used to provide a dataoutput to any follow-on digital processors that may be used in the EWapplication.

The single-chip receiver solution depicted in FIG. 5 completelysupplants an MCM embodiment of an EW receiver that, as discussed, doesnot satisfy the need for high integration levels. In the subject system,and referring to FIG. 6, the same RF pre-select/attenuation/notch unit44 feeds an indium phosphide heterojunction bipolar transistor HBTanalog-to-digital converter 82 coupled to a 64:1 demultiplexer 84 asillustrated. These two units correspond to a module 86 as illustrated.

The outputs of unit 44 are likewise supplied to a low band pass unit 86and a high band converter 88, the outputs of which are switched at 90 tothe aforementioned IF filter bank 58, with the high band converter beingprovided with a clock and local oscillator generator 92 having an outputcoupled to an analog-to-digital converter 94, with a digital automaticgain control 96 coupled to a variable-gain amplifier 98. The output ofanalog-to-digital converter 94 is coupled to a 16:1 demultiplexing unit100.

The demultiplexing units 84 and 100 provide respectively 128 outputs at625 MHz, with 256 pads of LVDS, but with a current draw of 5 watts forthe I/O that this driver represents.

For demultiplexer 100, there are 96 outputs at 625 MHz, with 192 padsLVDS but with a current draw of 4 watts of I/O, such that the combinedcurrent draw for these two demultiplexers is 7 watts. The I/O for a CMOSDSP 102 to which these two demultiplexers are coupled involves aninterface for demultiplexer 84 having 128 inputs at 625 MHz, 256 padsLVDS, which has an associated current draw of 2 watts of I/O. On theother hand, the output from demultiplexer 100, when interfaced to CMOSDSP 102, having 96 inputs at 625 MHz and 192 pads LVDS, has anassociated current draw of 2 watts. At this point, one has expended 13watts in the I/O process due to the drivers and demultiplexersassociated with the interfacing of the modules.

The output of the CMOS DSP is applied to a serializer 104 in a module106 that includes an additional demultiplexer 108. It is noted that theCMOS DSP output has a 128 I/O at 625 MHz, with 256 pads LVDS and has acurrent drain of 2 watts of I/O, bringing the total current draw of thismodular system up to 15 watts. Note that the internal demultiplexer 108in module 106, which offers a 14:1 demultiplex capability draws 1 watt,which in either the subject single-chip receiver-on-a-chip or the MCMversion thereof, is necessary.

What can be seen from the MCM version described above is that, due tothe interfacing of the various modules of the MCM receiver, one wastesapproximately 15 watts of the total 25-watt power consumption, whichwasted power is recouped by using the subject single-chip receiver.

Referring to FIG. 7, the difference between the standard modularapproach and the subject system is readily apparent.

In the prior art, electronics array 120 is serially coupled to antennaelements 122 to drive a low noise amplifier 124 through a silicon oxidecable 126 coupled to a RAIU 128, which is switched to the output of thecable via switching circuit 130. The output of switch 130 is filtered at132 and is applied to a down-conversion mixer 134 supplied with theoutput of a first local oscillator 136. The down-converted output isfiltered at 138 and is amplified at 140, whereupon it is applied via acable 142 to a second stage down-converter 144. This down-converter hasits own switching circuit 146 coupled to a filter 148 and to a mixer 150that serves as a down-converter, with the mixer being supplied with asecond local oscillator signal over line 152. The output of the mixer isapplied to a filter 154 and thence to an amplifier 156, the output ofwhich is applied to an analog-to-digital converter 158 coupled thence toa DSP 160.

Rather than using the two stages of down-conversion associated with theMCM version of the EW receiver and due to the availability of higherspeed analog-to-digital converters, analog-to-digital converter 158 withimproved technology is used in the single-chip receiver-on-a-chip IC 160version of an EW receiver. Each IC 160 is coupled to its own individualantenna element 162, the outputs of which are supplied to fiber opticdrivers 164, which drive fiber optic cables 166. These fiber opticcables are provided to an RF unit 168, which has a fiber optic receiverand switching unit 170, the output of which is applied to a digitalsignal processor 172.

What will be seen in this comparison is that a large number ofindividual modules associated with prior art EW receivers can beeliminated along with cabling between the modules due to the use ofsuper high-speed silicon-germanium architectures, which results in theadvantages listed in the Table of FIG. 8.

Referring now to FIG. 8, it can be seen that the operating frequency isextended down to 0.3 GHz, whereas the instantaneous bandwidth goes from800 MHz to 2,000 MHz, with the dynamic range increased from 50 dB to 60dB. Most importantly, the size is reduced from 200 cubic inches to 7cubic inches, with a rate reduction over the MCM version of ten poundsto one pound and an MTFB going from 1,000 hours to 10,000 hours.

More specifically, the present invention uses IBM's latest 7 HP and 8 HPSiGe HBT foundry process in which an A/D converter (4 bits@10 GHz) isused which is monolithically integrated with a 40 GB/s serial link.Other building blocks are also developed including a variable gainamplifier and an LNA.

I. Architecture

The frequency plan uses dual IF frequencies of 2-4 GHz and 6-8 GHz.Those IF frequencies map into the first and second Nyquist bands of theADC operating at 10 Gsps. These bands correspond to 2 of the RFfrequency bands that are needed, so those bands can be direct-sampledwith maximum spur-free dynamic range. For other frequency hands, thedual IF frequencies allows us to use 1 LO Frequency to get two RF bands,simply by selecting the appropriate IF alias-band frequency. Thisreduces the number of VCO frequencies that will be required to obtaincomplete band coverage. The single-conversion also minimizes the designcomplexity associated with off-chip interfaces that are needed fordouble-conversion.

SiGe technology provides a revolutionary benefit in wideband systems, asit enables large amount of digital processing to be done on-chip,without the latency and power dissipation that is incurred intraditional multi chip system implementations.

II. Components

1. Low Noise Amplifier

The wideband amplifier is implemented using a distributed amplifiertopology, which offers high gain, wide bandwidth, and low noise figure.Each stage of the distributed amplifier is implemented ascommon-emitter-common-base amplifier. For all the stages of distributedamplifier, an emitter degeneration resistor is included to reduce thelow frequency gain. A shunt capacitor is attached to the emitter toeliminate the gain reduction due to the degeneration resistors at highfrequency.

Referring now to FIG. 9, what is shown is a photograph of the fabricatedLNA chip. Connection 173 is a single-ended input referenced to multipleground connections 175. Five amplifier gain stages 181 amplify thesignal coupled off of the input and sum their respective outputstogether in a spatially distributed fashion. The output is then sent offof the chip at the single-ended port 177, which is also referenced tothe ground connections 175. The remaining connections 179 are connectedtogether on-chip and externally connected to the voltage bias supply,V_(EE).

DC biasing current and the number of stages in the LNA significantlyaffects the gain and noise performances. An increase in number of stagesand biasing current will increase both gain and noise figure, and viceversa. The optimum number of stages and single stage collector biasingcurrent is found to be five stages and 4 mA respectively. High frequencymeasurement of the LNA is carried out using on-wafer short, open, loadand thru (SOLT) calibration with an HP8510C vector network analyzer. Thenoise figure of the LNA is measured from 1 to 18 GHz using an HP 8971Cnoise figure meter, an ATN tuner and noise measurement software. TheS-parameter and noise figure measurement results are shown in FIG. 10.The 2-18 GHz LNA has a 15 dB gain, 5 dB noise figure, and VSWR betterthan 2.5:1 at both input and output ports. Two-tone third orderintercept point is also measured using a microwave source sweeper at18.0 GHz with 1 MHz frequency spacing. The measurement showed that thewideband LNA has an OIP3 of 15.5 dBm. In the next phase the noise figureis improved to 3 dB using the 8 HP process.

2. Variable Gain Amplifier/Mixer

The primary application of the VGA is automatic gain control in the IFpath. But the high linearity, wide range in gain control, and broadbandwidth also lends this circuit well to analog multipliers,down-converters, and up-converters.

Silicon germanium (SiGe) technology offers considerable advantages overother semiconductor technologies in this VGA application. SiGe HBTs have2-3× greater Ft than CMOS, enabling higher frequency, broader bandwidthcircuits. When using the IBM 8 HP technology, the bipolar transistordevices will exhibit Ft's of 220 GHz, while a leading 0.13 um RF-CMOSprocess has an Ft of only 80 GHz. SiGe does not offer any noise figureadvantages in LNA designs, but the high Ft allows large improvements inbandwidth.

The 1/f noise in SiGe HBT devices is considerably lower than CMOS. Thisallows much lower phase noise VCOs which can be used for on chip clocksources for A/D converters. The IBM7 HP 1/f corner is <1 kHz, while CMOSdevices exhibit 1/f corners greater than 1 MHz.

The VGA is based on a Gilbert cell topology utilizing a Cherry-Hooperbandwidth enhancement technique. Feedback and emitter degeneration areused to enhance linearity. FIG. 11 shows the micrograph of the IC andFIG. 12 shows a simplified schematic of the VGA core.

Referring now to FIG. 11, it will be appreciated that the VGA core 180is in essence a circuit to allow one to create and measure the interfaceparameters for the individual parts of the overall system one iscreating. By being able to measure the individual parts, one can thenadjust how the interfaces go together when one assembles the largercircuit.

VGA core 180 has a number of connection points that are in the form ofconnectors. At the left-hand side of the circuit are RF-in connectionpoints 182, which are surrounded by ground connectors 184. The RF-inconnection points are surrounded by a number of grounds to enhance theisolation of signals between the input-output and LO or gain controls.

The gain/LO-in connectors are illustrated at 186. These connectors aresimilarly surrounded by grounds 188. Note that there are a number ofspare connectors 190 that surround gain/LO inputs 186.

As illustrated at the right-hand side of the circuit, connectors 192correspond to the RF/IF-out signal connections, again similarlysurrounded by ground connectors 194.

At the bottom of this circuit are power connectors 196, attendantgrounds 198 and spares 200. Note that the actual active circuit isillustrated at 202, with the other elements 204 being power conditioningcircuits.

Referring now to FIG. 12, the circuit diagram corresponding to circuit180 of FIG. 11 assumes that it has a signal input corresponding toinputs 182 of FIG. 11, here illustrated at 201. These inputs go to thebases of the transistors of a transconduction input section 203 thatincludes NPN transistors 205 and 207. Here the input signals are appliedto the bases of these transistors. This transconduction input sectionconverts a voltage to a high-impedance current output, which is appliedto a four-quadrant gain control section 209 and more particularly to theemitters of transistors 211 and 213, as well as to the emitters ofback-to-back transistors 215 and 217.

The gain/LO inputs at 186 are control signals that are applied at 219and 221 to respective bases of transistors 211, 213, 215 and 217. Thissection creates a current-steering function that steers the differentialcurrent coming from the transconductance section as a function of thevoltage input on the inputs 219 and 221. That differential current isthen applied to a transimpedance output section 231 and moreparticularly to the bases of NPN transistors 233 and 235. These twotransistors form the transimpedance function of this circuit, whichconverts the current output from the gain control section that is at ahigh impedance, into a low-impedance input that drives the voltageoutput at terminals 237 and 239.

The top and bottom of the FIG. 12 circuit act as current generationsources for biasing the various impedance and gain-control sections ofthe rest of the circuit. The circuit elements for these biasing circuitsinclude transistors 241, 243, 245, 247, 249, 251 and 253, along withtransistor 255. In short, the aforementioned circuitry generates thecurrent that biases the circuit.

In operation, the voltage input signal is applied to differential inputports P1 and P2. The gain control section multiplies the input signal bythe gain voltage on differential ports P3 and P4. The signal passingthrough the gain control section is a current signal with minimalvoltage swing. This reduces the impact of parasitic capacitances andmaximizes bandwidth. The current signal is converted back to voltage inthe upper transimpedance section. The low input impedance ensures smallvoltage swings. This technique of impedance mismatching, where a lowinput impedance is driven by a high output impedance current sourceenhances bandwidth.

For testing the 1.28×2.00 mm chip was bonded to an alumina substrate. DCpower and control signals were wire bonded to the chip, while thehigh-speed ports used 40 GHz probe heads. External baluns where used tointerface the single-ended test equipment signals to the differentialports of the VGA.

The VGA was also tested as a down-converting mixer. Even though this SoCbuilding block was not optimized for this purpose, it performed verywell with a 26 GHz RF signal, 19 GHz LO signal, with a 7 GHz IF output.Note, the VGA presented here represents the baseline for the mixer cell.

A family of gain vs. frequency curve measurements is given in FIG. 13.This measurement was performed by wafer probing the chip on a substrate.The smooth red lines are simulation result, while the different coloredlines are 10 dB gain steps. Note that the gain is very flat at any gainsetting, a highly desirable feature of a VGA. The gain also changes in10 dB steps with 10 dB commanded gain steps. There are a couple ofslight deviations from simulation however.

2. A/D Converter and Serializer

One of the key issues in a high speed ADC is the problem of transferringlarge amounts of generated digital data from the ADC to the processingunit. In many applications ADC and the processing unit are notco-located. Transferring many lines of data in parallel is costly,unreliable, and prone to noise and electrical crosstalk. In the subjectinvention a novel approach is taken by using the existing fiber optictransceiver technology to multiplex the digital data into a higher ratestream and transferring it via a single fiber link. The main componentsof this chip are a 10 Gsps ADC and a 40 Gbps serializer. Additionalblocks such as digitally-generated dither are included on the chip toboost the performance of the ADC.

The 4-Bit 10-Gsps flash ADC was fabricated in IBM 7 HP SiGe technology.The ADC shows 45 dB of Spurious Free Dynamic Range. Digital output ofthe converter is multiplexed up into a single 40 Gbps stream. Internaldithering allows for improved spur performance. The converter uses a newwideband Track/Hold circuit and incorporates a novel layout in thequantizer section for speed improvement. FIG. 14 shows the top-levelblock diagram of the ADC/Serializer.

As can be seen in FIG. 14, the ADC/Serializer, here illustrated at 220,includes a timing control circuit 222, a clock select circuit 224, aclock multiplier unit 226 (CMU), all of which are used in controlling anA/D 4-bit analog-to-digital converter 228 coupled to a demultiplexer230, in turn coupled to a 16-bit parallel scrambler/pseudo-random writesequencer 232, in turn coupled to a 16:1 multiplexing circuit 234. Theoutput of multiplexing circuit 234 is amplified at 236 to provide a40-GHz serial bit stream of data at outputs 228.

At the bottom of this figure, additional control signals come in atconnectors 230, which are applied respectively to a parallelpseudo-random write sequence generator 232 coupled to a 4-bitdigital-to-analog converter 234, in turn coupled to a low pass filter236 having its outputs amplified at 238 and added to the inputs 240 toanalog-to-digital converter 228.

Note that what constitutes the input signal at 240 is the conditioned RFsignal from the variable gain amplifier/mixer described in FIG. 12.

Note also that a control signal on terminal 242 is used to enable thedigital output on the serial data link 228. Moreover, a 4-bitdigital-to-analog converter 244 is coupled to the output ofanalog-to-digital converter 228 controlled by a signal at 246, with theoutput of digital-to-analog converter 244 constituting a digitalreproduction of the input signal that can then be used as an outputtransponder response. This output is available at 248.

It will be appreciated that the inputs at 250 to timing-and-control unit222 are detail controls for the RF conversion and data formats that willbe used within the circuits, in addition to threshold control signalsfor determining response levels.

Also note that inputs 252 to CMU 226 are used for inputting a referenceclock signal for the internal clock generating function of the system aswell as the specific controls for the particular frequency of the sampleclock that will be generated.

It will also be noted that outputs 254 constitute test outputs to verifythat the internal clock generator function is working properly.

In operation, a dither generation circuit adds a controllable lowpassdither to the input of the ADC. This dither helps reduce spurs in thefrequency band of interest. The dither generation circuit consists of a4-bit pseudo-random write sequence generator, a 4-bit digital-to-analogconverter and a lowpass filter.

After the encoding section the 4-Bit binary code with data rate of 10Gbps is fed to a demultiplexer. This demultiplexer reduces the data ratedown to 2.5 Gbps suitable for the scrambler section. The output of thescrambler section drives a 16:1 multiplexer. This multiplexer uses aquadrature 20 GHz clock to generate a full 40 Gbps data output. The 20GHz clock is generated by an internal clock management unit (CMU), whichaccepts an external reference frequency of 625 MHz. The main challengein the final multiplexer/driver is the quality of the eye diagram and agood output return loss. These issues have been addressed by variouscircuit and layout techniques. The output 40 Gbps stream has ˜200 fsecrms jitter and ˜800 mVpp differential swing. An on chip DAC is includedto aid in characterizing the ADC performance.

FIG. 15 shows the spectrum of the DAC output with and without dithering.Reduction of spurs is noticeable as a result of the dithering. Theremaining spurs mainly are due to the limited Spur-Free Dynamic Range(SFDR) of the DAC. Note, the ADC performance can be improved to 6 bitswith 60 dB of dynamic range.

3. Ultra Wideband ADC

A potential architecture enhancement to the subject receiver-on-a-chipincludes the addition of a parallel receive path which monitors a verywide bandwidth so as to maximize the probability of intercept of highpriority threats. A 3-Bit 20-Gsps ADC is the critical component of thedesign providing up to 8 GHz of observable bandwidth. This part achievesSpur-Free Dynamic Range of 22 dB in the first Nyquist zone and SFDR of16 dB in the second Nyquist zone. The converter uses a new widebandTrack/Hold circuit and incorporates a novel layout in the quantizersection for speed improvement. Power consumption of the ADC core is ˜0.6W and total power consumption of the chip including the 50 Ohms CMLdrivers is ˜3 W. The IBM SiGe 8T process was used. Note that it ispossible to increase the speed of this ADC to over 40 GSPs.

FIG. 16 shows a top-level block diagram of the ADC/DAC. As can be seenin FIG. 16, the ADC/DAC includes a data input 260 coupled through anamplifier 262 to an analog-to-digital converter 264 clocked by a 20-GHzclocking signal 266 amplified at 268 and applied to analog-to-digitalconverter 264 on line 270.

The output of analog-to-digital converter 264 is applied to ademultiplexing circuit 272 and also to a digital-to-analog converter274.

Buffers 276, 278 and 280 are respectively coupled to the outputs ofdemultiplexer 272, a divide-by-eight circuit 282 and a digital-to-analogconverter 274 to provide a digital data stream with clocking, at leastinsofar as the outputs of buffers 276 and 278. The output of buffer 280constitutes an analog RF output used for a transmit jamming waveform.

In operation, the differential input is fed to an input buffer thatdrives the analog-to-digital converter section. The analog-to-digitalconverter consists of Track/Hold, circuit, quantizer section,bubble-correction circuitry and binary encoding. One of the criticalsections of a high-speed analog-to-digital converter is the Track/Hold(T/H) section. In this design a T/H circuit with over 65 dB SFDR isimplemented. The T/H uses two sets of current switches: “Main” currentswitches function as in conventional track-and-hold circuits. Additional“helping” current switches allow optimization of switching speed whileproviding proper reverse biasing during the hold mode time.

After the encoding section the 3-Bit binary code with data rate of 20Gbps feeds two sections: first the Internal DAC and second theDemultiplexer. Using a high-speed DAC to convert the digital data toanalog waveform is a very convenient way of testing and characterizingthe key features of the ADC. For 3-Bit operation, the DAC linearity andspeed are good enough not to degrade the observed ADC performance.

FIG. 17 shows the measured spectrum of a second Nyquist zone input(19.02 GHz). As can be seen, a spur-free dynamic range of 26 dB ismeasured. This is in very close agreement with the simulations. Thisperformance makes direct sampling of RF signals as high as 20 GHz aviable solution.

The Demultiplexer section is utilized to reduce the data rate of the3-Bit binary code from 20 Gbps to 2.5 Gbps. This requires demultiplexingby a factor of 8, resulting in 24 lines of data at 2.5 Gbps. Asynchronous clock at 2.5 GHz is also provided to facilitate externaldata acquisition.

Throughout the design special attention has been paid to the timingbetween different blocks. To reduce aperture jitter, buffers and gainblocks are included in the clock section to make the clock as square aspossible. The power supply for the output drivers is chosen to be 1.8vto facilitate easy interface with CMOS chips.

Those skilled in the art will appreciate that a novel way to architectnew Electronic Warfare systems has been described. An LNA, VGA, and anintegrated ADC/Serializer are combined and all worked to the programspecifications. The IBM SiGe BiCMOS process provides high RF performanceusing HBTs along with high levels of digital integration using CMOS.This is essential for wideband circuits where transferring signals fromchip to chip is costly in terms of power and I/O and reliability.

While the present invention has been described in connection with thepreferred embodiments of the various figures, it is to be understoodthat other similar embodiments may be used or modifications andadditions may be made to the described embodiment for performing thesame function of the present invention without deviating therefrom.Therefore, the present invention should not be limited to any singleembodiment, but rather construed in breadth and scope in accordance withthe recitation of the appended claims.

What is claimed is:
 1. An electronic warfare receiver system light andsmall enough to be located on an unmanned aerial vehicle and having apower consumption minimized to maximize endurance, comprising: an arrayof antenna elements on said unmanned aerial vehicle; a lightweightmonolithic broadband receiver-on-a-chip located at each antenna element;fiber optic cables coupled at one end to the output of respectivereceivers-on-a-chip; and, a processor coupled to the other ends of saidfiber optic cables for processing the individual outputs of saidreceiver-on-a-chip, wherein each of said receivers-on-a-chip has avolume of less than 7 cubic inches, and each of said receivers-on-a-chipweighs less than a pound.
 2. The system of claim 1, wherein there are noI/O drivers between the circuits on said receiver-on-a-chip, whereby thepower consumption of said receiver-on-a-chip can be made less than 10watts.
 3. The system of claim 1, wherein said broadbandreceiver-on-a-chip manufactured in a monolithic process usingsilicon-germanium transistor technology so as to produce transistorswitching speeds in excess of 100 GHz to effectuate a 0.03 GHz to 18 GHzoperational bandwidth.
 4. The system of claim 3, wherein saidreceiver-on-a-chip includes super high-speed analog-to-digitalconverters running at over 100 GHz to reduce the requirement for numbersof IF stages including down-converters and local oscillators.